Electronic arc welding station with input voltage compensation

ABSTRACT

An improved welding system is shown. A single high capacity power supply (10) provides operating power for several weld selector stations (16). Each weld selector station (16) operates independently, is adjustable, and allows a welder to obtain the voltage and current characteristics desired for his particular welding operation. Each weld selector station (16) operates over a wide range of input voltages so that long runs of low-voltage interconnecting cable (14,15) can be used without adversely affecting the quality of the weld. Furthermore, the weld selector station (16) is small, lightweight and easily transportable. The result is a more efficient, less expensive, versatile welding system.

This is a continuation application of U.S. patent application Ser. No.118,613, filed Nov. 9, 1987, now U.S. Pat. No. 4,887,046 which is adivisional application of U.S. patent application Ser. No. 791,224,filed Oct. 25, 1985, and is now U.S. Pat. No. 4,716,274, dated Dec. 29,1987.

TECHNICAL FIELD

The present invention relates to welding power supplies and inparticular to improved distributed welding power supply systems having asingle, relatively high voltage power source, and a plurality ofphysically distributed power regulators and DC-to-DC converters for useby a plurality of individual welders.

BACKGROUND OF THE INVENTION

The present invention is an improvement to welding power suppliesparticularly suited for environments in which a relatively large numberof welders are working in the area. The present invention isparticularly useful in environments in which welding must be done inrelatively small cramped areas. In particular, the environment of ashipyard is one in which the present invention is particularly useful.

As is known to those skilled in the art, welding jobs in shipyardsnormally include a relatively large number of welders weldingsimultaneously. Additionally, in the construction or repair of a ship,it is often necessary for a welder to work in relatively crampedquarters within the interior of a ship.

Prior art arrangements have normally included a relatively large numberof power supplies, one for each welder. Such an arrangement is timeconsuming and expensive since it requires multiple drops of three-phasehigh voltage AC lines to supply the various power supplies. As is knownto those skilled in the art, accidental cutting of high voltage AC powerlines leads to much more dangerous situations than similar accidentsinvolving the lower voltage DC lines.

Furthermore, lack of reasonable access to the AC power lines and/or theimpracticality of or undesirability of moving the prior art power supplysometimes creates a situation wherein the prior art power supply issubstantially removed from the welder and a long run of welding cable isrequired. Since the low voltage high current outputs of such weldingpower supplies require large gauge cables, pulling long runs of thesecables around a shipyard or similar environment is cumbersome, resultsin increased I² R losses as the cable length increases, and causesvariations in the quality of the weld because of the distributedinductance, capacitance and resistance of long runs of cable.

Additionally, the use of multiple individual welding power suppliesnecessarily means duplication of one of the most expensive components ofa power supply, a large, high current three-phase transformer.

Thus, as explained in greater detail hereinbelow, the present inventionprovides a distributed welding power supply system which overcomes someof these disadvantages. The system of the present invention generallyconsists of a single high power regulated power supply and a pluralityof distributed weld selector stations. Within each weld selector stationthere are a number of elements essentially similar to a smaller weldingpower supply. Therefore, it is appropriate to consider other aspects ofthe background of the art of solid state welding power supplies inconnection with the weld selector stations used in the present system.

Again considering the environment of a shipyard as a typical environmentfor welding, it is highly desirable to provide a power supply which isusable in a relatively large number of welding processes. In particular,there are a number of variations in the electrical outputcharacteristics of a welding power supply which affect its utility inparticular welding processes. Among these are the output impedance andthe turn-on and turn-off time or the dV/dt (dI/dt) characteristics ofthe voltage output (current output) of the supply. For example, intungsten inert gas (TIG) and stick welding, it is generally desirable tohave a power supply with a high output impedance so that itscharacteristics approximate a constant current source. In metal inertgas (MIG) and pulse arc welding, it is often desirable to have a powersupply with a low output impedance which approximates a constant voltagesource. There are, of course, other types of welding which require acompromise between these characteristics.

Another aspect of a welding environment well known to those skilled inthe art is the fact that large transients are present in the maincurrent carrying cables within a welding system. The use of pulse widthmodulators to control solid state switching devices to adjust outputvoltage has been known in the art for some period of time. Generally,such pulse width modulators are constructed using relatively low voltagesolid state integrated circuits. Such circuits are provided with wellregulated and well by-passed low voltage power supplies so that theywill operate properly.

One problem with prior art power supplies of this type occurs fromtransients which occur on the DC input to a welding power supplyswitching regulator operated by the pulse switch modulator. When an arcis struck, there is normally a large current surge initially drawn fromthe output of the power supply which normally lowers the voltage at theinput of the regulator. A problem which arose in designing a system ofthe present invention was the fact that multiple welders are operatingweld selector stations off of a single power supply output. As variouswelders using the system strike arcs, the overall output voltage of themain supply, and thus the input voltage to the various weld selectorstations, will drop. Thus, it is desirable to provide a feedback systemfor pulse width modulators used in a distributed welding power supplysystem which will respond to changes in the input voltage to the weldselector station.

Additionally, in some welding environments there are a large number ofremote welding units which include wire feed motors and shielding gasvalves, for example, of the type shown in applicant's U.S. Pat. No.4,119,830, issued Oct. 10, 1978. The voltage supply to the wire feedmotor normally fluctuates with the voltage supplied by the power supplyoutput. Also, in some types of MIG welding, the output voltage is lowerthan that required to drive the wire feed motor. Thus, there is a needto assure that the voltage to the wire feed motor will not becomeexcessive so as to damage the motor or inadequate so as to causestalling or erratic operation.

Furthermore, in many prior art systems, a high power resistance box isused to obtain the desired output voltage, current and impedance. Theseboxes can consume substantial amounts of power which must be dissipatedas waste heat, lowers the energy and cost efficiency of the system, andrequires additional air flow or cooling.

Lastly, there is a problem in the state of the prior art, not limited tothe environment of welding power supplies, which has been overcome bythe present invention. As is well known to those skilled in the art, thedesign of reliable push-pull amplifiers, whether they be linearamplifiers or push-pull devices used in switching power supplyregulators, have always been considered to require closely matched pairsof output devices. This goes back to the days of push-pull vacuum tubeaudio amplifiers. In the design of push-pull solid state amplifiers,this problem has become more critical since excessive gain in onetransistor of a push-pull amplifier stage normally leads to greaterpower dissipation and thus a higher operating temperature for thatparticular transistor. As the operating temperature increases, the betaof the transistor tends to increase, and this ultimately leads to acondition known as thermal run-away in which the higher beta transistorwill be destroyed.

As is also known to those skilled in the art, matched pairs oftransistors of a particular type tend to be considerably more expensivethan individual transistors of the same type. This cost increase becomeseven greater when relatively high current devices, such as those used inrelatively large switching power supplies, are used. Thus, there is aneed in the art to provide a practical and reliable arrangement fordesigning push-pull amplifier output stages (which may be used inswitching power supplies, linear amplifiers, and other applications forpush-pull topology) which can eliminate the requirement for transistorshaving closely matched betas and thermal characteristics.

SUMMARY OF THE INVENTION

The present invention overcomes the drawbacks of the prior art recitedabove and provides some less expensive advantageous arrangements foraccomplishing some end results known in the prior art. Broadly stated,the system of the present invention is a distributed welding powersupply system in which one large relatively high voltage high powersupply (80 volts at 1500 amps continuous in the preferred embodiment) isused to power a plurality of relatively small weld selector stationunits which are essentially high current DC to DC converters. Each weldselector station may be operated in a constant current mode, a constantvoltage mode, or a variable combination of the two. Additionally, eachweld selector station uses a pulse width modulated switching to controlthe output power from the weld selector station. The pulse widthmodulator controlling the output of each weld selector station isarranged to respond to drops in the input voltage rather than use ofconventional arc voltage feedback in order to maintain a nominallyconstant output.

Because each weld selector station unit is supplied from a highervoltage source than most conventional welding equipment, the weldselector station may be connected the main power supply via smallergauge cabling than that normally used to connect a weld torch to aconventional 40-volt supply. The weld selector stations of the preferredembodiment are such that they can be readily implemented in a smallcabinet, weighing approximately eighty pounds, which may be easilytransported into confined spaces by two people.

Among the advantageous features of the weld selector station of thepresent invention are a switch for automatically selecting a source ofinput voltage for a wire feed motor to be used with a welding torchconnected to the output of the weld selector station. For applicationsin which the nominal output voltage from the weld selector station isadequate to drive the wire feed motor, the wire feed motor circuitry isconnected directly to the output of the weld selector station. When alower output voltage is selected, for example in a number of metal inertgas (MIG) processes, the system automatically switches the source of thewire feed motor voltage to one of several user selectable voltagedividers, one of which is selected to pick off a predetermined portionof the input voltage to the weld selector station.

Another advantageous aspect of the weld selector station of the presentsystem, which applies to welding power suppliees in general, is thedrive circuitry for the main switching transistors. The base of theswitching transistors are driven by a pair of metal oxide semiconductorfield effect transistors (MOSFETs), one of which is a P-channel type andone of which is an N-channel type. The FETs have their source terminalstied together to the base of the main switching transistor. The gates ofeach of the FETs have independently adjustable impedances connectedbetween them and the source of a pulse width modulated control signalfor controlling the output of the weld selector station. By adoptingthis arrangement, the independently adjustable gate impedances variesthe rise and fall times of the output of the main switching transistors.As is known to those skilled in the art, various welding processesrespond differently to the transient characteristics of the output fromthe switching device from a switching power supply.

Additionally, the P-channel FET is selectably switchable in and out ofthe base drive circuit. When it is in the circuit, it causes the base ofthe main switching transistor to be driven to a voltage lower than theemitter, thus depleting charge carriers in the base region, therebycausing the main transistor to turn off very rapidly. When more gradualturn-off the switching transistors is desired, the drive to theP-channel FET may be reduced or it may be removed from the base drivecircuit.

Additionally, the present invention includes a novel arrangement forbiasing a pair of transistors used in a push-pull configuration. In thepreferred embodiment of the present invention, the push-pull stage inquestion is driving the primary of a power transformer which is used ingenerating well regulated lower voltage power supplies used to power anumber of the lower voltage integrated circuit components in thepreferred embodiment. However, the principle of this aspect of thepresent invention has utility in a large number of other applications inwhich power transistors are interconnected in a fashion in which theyneed to conduct substantially equal amounts of average current in orderto avoid the circumstances of one transistor going into thermalrun-away. For example, push-pull power output stages in audio amplifiersmay benefit from this aspect of the present invention. Also, parallelpower transistor configurations may be rendered less expensive byadoption of this aspect of the present invention.

Generally stated, this aspect of the present invention can be used toprovide considerable cost savings to the manufacturer of devices whichnormally require matched pairs of set of transistors. As is known tothose skilled in the art, matched pairs of transistors are ones (oftenobtained from the same die during fabrication) which have closelymatched characteristics. Of particular concern is the forward currentgain or beta, of transistors having similar or common base drives. Asnoted above, in the preferred embodiment, the arrangement meeting thisrequirement is that of a push-pull switching amplifier driving aninductive load.

As is known to those skilled in the art, if one of the transistors insuch an arrangement has a somewhat higher beta than the other. It willdraw more current than the other transistor. This leads to a situationin which the junction temperature of the higher beta transistor becomeshigher than that of the other transistor. As junction temperatureincreases, beta increases and thus the situation is exacerbated by aform of positive feedback.

As is known to those skilled in the art, eventually this situation leadsto thermal run-away and destruction of the higher beta transistor. Thisis why such amplifier arrangements are conventionally constructed usingmatched pairs of devices. In the situation of parallel drivetransistors, unless power wasting emitter resistors are used, onetransistor tends to "hog" more current than the others. Under thesecircumstances, the collector-emitter saturation voltage is alsodecreased as the transistor heats up, and destruction of one of thedevices often results, frequently followed by a chain reactiondestruction of the other transistors as they try to accommodate theincrease in current caused by the destruction of the first transistor.

The main drawback to matched transistors, particularly relatively highpower devices, is that they are much more expensive than buying aplurality of transistors of the same type which do not necessarily havetheir characteristics closely matched, other than the extent to whichthey are matched by virtue of being the same type device. The presentinvention provides a mechanism for using non-matched pairs oftransistors in circumstances in which the conventional wisdom of theprior art dictates that matched pairs be used.

Broadly stated, this aspect of the present invention provides thermallysensitive base drive to each of the transistors in question. Each of thethermally sensitive base drive elements is thermally coupled to a heatsink connected to the other transistor of the pair. In the preferredembodiment, each transistor includes a base emitter resistor having apositive temperature coefficient which is mounted on the heat sink orcase of the other transistor. As a second transistor becomes hotter thana first transistor, the base emitter resistor of the first transistorheats up, thus increasing its value. This has the effect of shuntingless of the input current from the base of the first transistor, thusincreasing the first transistor's base drive. Therefore, the firsttransistor begins conducting more current and the pair moves toward anequilibrium condition.

In the preferred embodiment, a potentiometer is connected as a variableresistor between the common connection between the emitters of the drivetransistors and a switch. The switch is configured to selectivelyconnect the other end of the variable resistor to the base of either ofthe drive transistors. This arrangement is used to initially calibratethe transistor pair. The switch is moved back and forth between the twobases as the variable resistance is adjusted until an initial conditionof equality between the currents conducted by the two transistors isestablished. Once the device is subsequently operated, theabove-referenced thermal feedback connection is used to maintain thisinitially established equilibrium.

Therefore, it is an object of the present invention to provide animproved system for providing welding power in an environment in which alarge number of welders are working simultaneously.

It is a further object of the present invention to provide a distributedwelding power supply system employing one master power supply and aplurality of lightweight weld selector stations which is less expensiveand more energy efficient than provision of an equivalent amount ofpower through the use of individual supplies.

It is a further object of the present invention to provide a switchingpower supply output stage for a welding power supply employing switchingtransistors which have selectively variable turn-on and turn-offcharacteristics.

It is a further object of the present invention to provide a pulse widthmodulated control switching welding power supply which has a selectivelyvariable pulse width which responds to a drop in the input voltage tothe regulator to increase the pulse width, thus maintaining a nominallyconstant output power.

It is also an object of the present invention to provide a variablevoltage output welding power supply which includes apparatus forautomatically switching the source of voltage used to drive a wire feedmotor at a remote welding torch.

It is still a further object of the present invention to provide aninexpensive implementation of a standard transistor circuit by providingapparatus which allows the use of unmatched pairs of transistors in acircuit configuration normally requiring matched pairs.

It is a further object of the present invention to provide a distributedwelding power supply system which allows a plurality of welders toselect different output voltages and use different welding methods suchas stick, TIG or MIG, while all being powered from a common powersupply.

That the present invention accomplishes these objectives, and overcomesthe drawbacks of the prior art noted above, will be apparent from thedetailed description of the preferred embodiment below.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of the preferred embodiment of the presentinvention.

FIG. 2 is a block diagram of the weld selector station of the preferredembodiment.

FIG. 3 is a schematic diagram of the switching and shutdown driver ofthe preferred embodiment.

FIG. 4 is a schematic diagram of the current detection circuits of thepreferred embodiment.

FIG. 5 is a schematic diagram of the STICK & TIG control circuit of thepreferred embodiment.

FIG. 6 is a schematic diagram of the low frequency pulsewidth modulatorof the preferred embodiment.

FIG. 7 is a schematic diagram of the high frequency pulsewidth modulatorof the preferred embodiment.

FIG. 8 is a schematic diagram of the high/low feeder voltage switchovercontrol circuit of the preferred embodiment.

FIG. 9 is a schematic diagram of the power supply circuits of thepreferred embodiment.

DETAILED DESCRIPTION

Turning now to the drawings, in which like numerals reference likeelements throughout the several drawings, the preferred embodiment ofthe present invention will be described. As shown in FIG. 1, thepreferred form of the present invention is a central welding powersupply 10 feeding a number of remote weld selector stations 16a-16n.Central welding power supply 10 contains a single AC-to-DC converter 11for providing power to the weld selector stations 16a-n.

Converter 11 is connected to a source of three-phase AC power (notshown), normally of 460 or 230 volts, by conductors 5, 6 and 7. Thepositive output of AC-to-DC converter 11 is connected by conductor 12 toa number of positive output studs 8a-8n. The negative, or return, outputof converter 11 is connected by conductor 13 to a number of groundedoutput studs 9a-9n. It will be appreciated that the studs 8 and 9 areconventional in nature and provide the connection point at the weldselector station for the welder.

Converter 11 provides an output of 80 volts DC (nominal) at a currentsufficient to power the desired number of remote weld selector stations16a-16n. In the preferred embodiment, converter 11 provides 1500 amps(continuous) current. Methods of construction of central power supply 10are well known to those skilled in the art. Remote weld selectorstations 16a-16n are connected by conductors 14a-14n and 15a-15n tostuds 8a-8n and 9a-9n, respectively.

Each remote weld selector station 16 can accept an input voltage of 30to 150 volts (80 volts nominal) and has its own controls for varyingvoltage and current characteristics. This allows a welder using a remoteweld selector station 16a to adjust the voltage and current outputcharacteristics to match the type of welding that welder is performingwithout affecting the voltage and current characteristics of remote weldselector stations 16b-16n.

It will be appreciated the present invention requires only a singlecentral power supply 10 and a single AC power connection instead of themultiple power supplies and AC power connections of conventionalsystems. Furthermore, it will be appreciated that since the remote weldselector station 16 weighs only 80 pounds it can be readily moved fromplace to place as required.

It will also be appreciated that, in conventional welding systems,because of the high currents involved, the length of the electricalcable between power supply and welder has a significant effect on thequality of the welding performed. Each remote weld selector station 16therefore acts as a buffer between the central power supply 10 and thewelder. It will therefore be appreciated that the present inventioneffectively reduces the length of electrical cable to the short distancebetween the remote weld selector station 16 and the welder. The presentinvention therefore reduces the cost of a welding system by minimizingthe amount of expensive control cable and gas hose used. The presentinvention also reduces the cost of a welding system by consolidatingseveral smaller power supplies into one central power supply 10 andgreatly reducing the number of AC power connections that must beinstalled.

Weld selector station 16 may be briefly described as a high efficiency,overload protected, selectable constant current or constant voltageswitching power supply with automatic compensation for input voltagefluctuation, and selectable low frequency pulse, high frequency pulse,and chopped low frequency pulse outputs.

Turn now to FIG. 2, which is a block diagram of a remote weld selectorstation 16. High efficiency is obtained by the use of switchingtransistors 56a-56f, inductor 62, and a free wheeling diode 61.Transistors 56a-56f are rapidly switched on and off to provide a desiredcurrent or voltage at output studs 17 and 18 instead of the conventionalmanner of using an array of high power dissipation resistors.

The positive 80 volt DC conductor 14 is connected to the anode of adiode 20. Diode 20 protects weld selector 16 in the event that anegative voltage is applied to conductor 14. The cathode of diode 20 isconnected by conductor 21, the internal positive 80 volt DC line, to aresistance bar 25, the STICK & TIG control circuit 30, the low frequencypulsewidth modulator 32, the high frequency pulsewidth modulator 34, thecircuit power supplies 38, the high/low voltage switchover circuit 86,the positive terminal of 10,000 microfarad storage capacitor 22, and toone end of the coil of relay 43.

Storage capacitor 22 stores power from and absorbs voltage surges on VINconductor 14 and provides a low impedance source which can provide largewelding currents for a short period. Diode 20 also prevents capacitor 22from discharging into VIN conductor 14 in the event that the voltage onVIN conductor 14 should fall below the voltage on conductor 21. Diode 20and capacitor 22 therefore isolate the output of weld selector station16 from transients caused by other welding operations and from thedistributed inductance/capacitance/resistance characteristics ofconductors 14 and 15.

The ground connector 15 is connected to one end of a fuse 24 and to theground output stud 19. The other end of fuse 24 is connected byconductor 23 to the negative terminal of capacitor 22, the circuit powersupplies 38, the anode of diode 61, one end of ON/OFF power switch 46,and the high/low voltage switchover circuit 86. The other end of switch46 is connected by conductor 47 to the other end of the coil of relay43. Although only a single ON/OFF switch 46 is shown, it will beappreciated that it is representative of multiple, interlocking switchesso that relay 43 may be energized/de-energized from a local or remotelocation, such as an ON/OFF switch on the welding torch.

The other end of resistance bar 25 is connected by conductor 42 to onecontact of a normally open pair of contacts in relay 43. The othercontact of relay 43 is connected by conductor 44 to four snubbercircuits 50a-50d, six power transistor circuits 55a-55f, andarc-sustaining resistor 45. The other end of the snubber circuits50a-50d, of the power transistor circuits 55a-55f, and of arc-sustainingresistor 45 arc connected by conductor 41 to the cathode of diode 61,one end of inductor 62, and the return output of shutdown driver 37. Theother end of inductor 62 is connected by conductor 63 to one end ofresistor 64 and one input of overcurrent detector circuit 73. The otherend of resistor 64 is connected by conductor 65 to the other input ofovercurrent detector circuit 73, one end of resistor 66, and the MIG &PULSE stud 18. The other end of resistor 66 is connected by conductor 67to one end of inductor 70. The other end of inductor 70 is connected byconductor 71 to the STICK & TIG stud 17, one input of the high/lowvoltage switchover circuit 86, one end of 44 ohm resistor 93 and theinput of short circuit detector circuit 72. Inductor 62 and resistor 64provide regulation and smoothing of the output for MIG & PULSE welding.Inductor 70 and resistor 66 provide the additional output smoothingrequired for STICK & TIG welding. The other end of resistor 93 isconnected to 80 v return conductor 23 which is the ground for powersupply 11 (FIG. 1). Resistor 93 has two tap points which are connectedby conductors 91 and 94 to two of the inputs of high/low voltageswitchover circuit 86. Resistor 93 provides a minimum, stabilizing loadfor transistors 56a-56f when the weld selector station 16 is on but thewelder has not yet struck an arc.

Resistance bar 25 has several tap points. The first and second tappoints are connected by conductors 26 and 27 to the positive andnegative inputs, respectively, of STICK & TIG control circuit 30. Thethird and fourth tap points are connected by conductor 76 and 77 to thepositive and negative inputs, respectively, of peak current detectorcircuit 80.

The output of STICK & TIG control circuit 30 is connected by conductor31 to one input of high frequency pulsewidth modulator 34. The output oflow frequency pulsewidth modulator 34 is connected by conductor 33 tothe other input of high frequency pulsewidth modulator 34. The positiveand negative outputs of high frequency pulsewidth modulator 34 areconnected by conductors 35 and 36 to a first positive and a firstnegative input, respectively, of switching, shutdown and driver circuit37.

The outputs of overcurrent detector circuit 73 and short circuitdetector 72 are connected by conductor 74 to one input of peak currentdetector 80. The output of peak current detector 80 is connected byconductor 81 to a second positive input of switching, shutdown anddriver circuit 37. The common reference of overcurrent detector 73,short circuit detector 72, and peak current detector 80 are connected byconductor 75 to the second negative input of switching, shutdown anddriver circuit 37.

The driver output and the reference output of switching, shutdown anddriver circuit 37 are connected by conductors 40 and 41 to the driverinput and reference input of power transistor circuits 55a-55f,respectively.

The circuit power supplies 38 provide the following output voltage toother circuits of the weld selector station: ±14, ±7, ±15, ±66.7, ±1.0.

Arc-sustaining resistor 45 has one tap point which is connected byconductor 82 to one input of high/low voltage switchover circuit 86.

One end and the wiper of potentiometer 84 are connected by conductor 83to a second input of high/low voltage switchover circuit 86. The otherend of potentiometer 84 is connected by conductor 85 to a third input ofswitchover circuit 86. The speed control outputs of switchover circuit86 are connected by conductors 95 and 96 to the speed control inputs offeeder control 99. The feeder power outputs of switchover circuit 86 areconnected by conductors 95 and 96 to the feeder power inputs of feedercontrol 99.

Power transistor circuits 55a-55f are identical and each contain atransistor 56 and an emitter resistor 57. The collector of transistor 56is connected to conductor 44. The emitter of transistor 56a is connectedto conductor 41 through resistor 57a. The base of transistor 56a isconnected to conductor 40.

Power transistors 56a-56f are each rated at 350 volts and 100 ampscontinuous. Although six NPN transistors 56a-56f are shown, it will beappreciated that the number of transistors 56 required will be dependentupon the maximum weld current desired and the ratings of the individualtransistor. It will also be appreciated that PNP transistors, fieldeffect transistors (FET), or other semiconductor devices can also beused. At present, bipolar transistors are preferred because of theirhigher ratings and lower cost.

Each power transistor 56 has an emitter resistor 57 valued at 0.55 ohms.It will be appreciated that the purpose of resistor 57 is to preventcurrent-hogging and destructive thermal runaway by any of the paralleltransistors 56. It will also be appreciated that FET's have a positiveresistance-temperature coefficient and do not suffer from the same typeof current hogging. Therefore, emitter resistors 57 would not berequired if transistors 57 were an FET. However, the performance ofparalleled FET's can suffer somewhat due to slightly differing gatecharacteristics. Therefore, if FET's are used for transistor 57 aresistor should be placed in series with the gate of each FET. The valueof the resistor will be dependent upon the gate characteristics of theparticular FET used.

The four snubber circuits 50a-50d are identical. Snubber 50a contains an8 microfarad capacitor 51a with one end connected by conductor 52a tothe anode of a fast recovery diode 53a and one end of a 0.25 ohmresistor 54a. The other end of capacitor 51a is connected to conductor44. The cathode of diode 53a and the other end of resistor 54a areconnected to conductor 41. Snubbers 50a-50d protect power transistors56a-56f from switching transients. It should be noted that the valuesfor capacitor 51a and resistor 54a are not the values that would beobtained by the use of standard snubber circuit design equations andtables because of the presence of arc-sustaining resistor 45.

As will be explained in detail below, power transistors 56a-56f areswitched on and off at a rapid rate. It will therefore be appreciatedthat when transistors 56a-56f are off, there will be no current tosustain the welding arc and the arc will be quenched. This undesirablecondition is prevented by arc-sustaining resistor 45 connected betweenconductors 44 and 41. Resistor 45 has a value of 2.4 ohms, which willpass sufficient current to keep the welding arc alive for a short periodwhile transistors 56a-56f are off. When transistors 56a-56f are on, thevoltage across resistor 45 will be reduced to a low value and thecurrent supplied by resistor 45 will be negligible compared to thecurrent conducted through transistors 56a-56f.

Arc sustaining resistor 45 also provides an additional benefit. Assumefor a moment that arc sustaining resistor 45 is not present, and thattransistor 56a develops a short. When driver 37 turns off transistors56b-56f, the full load current will flow through transistor 56a whichwill, in most cases, cause it to burn out and permanently become an opencircuit. When driver 37 turns transistors 56b-56f on again, they willhave to also pass the current that should have been passed by transistor56a. This extra current may cause transistor 56b to fail shorted, thenopened, and so on in a chain reaction until all six transistors havefailed in the open mode.

Now insert resistor 45. If transistor 56a develops a short, then whendriver 37 turns off transistors 56b-56f, the load current will bepartially absorbed by resistor 45. This prevents transistor 56a fromopening and therefore prevents the chain reaction from occurring.Resistor 45 thus causes shorted transistor 56a to stay shorted, insteadof opening, and preserves transistors 56b-56f.

Of course, if transistor 56a shorts, the weld selector station 16 willnot operate properly, but the cost of repair will be reduced.

Assume that transistors 56a-56f are on, that the welding equipment isconnected to the STICK & TIG stud 17, and that welding is in progress sothat current is flowing through inductors 62 and 70 and resistors 64 and66. It will be appreciated that inductors 62 and 70 will oppose anyrapid change in current.

Free wheeling diode 61, conductor 23, and fuse 24 therefore provide apath so that, when transistors 56a-56f turn off, the current throughinductors 62 and 70 will decrease gradually instead of instantaneously,thereby preventing a large negative voltage from appearing on theemitter of transistors 56a-56f. The decay rate of the current will bedetermined primarily by the values of inductors 62 and 70, resistors 64and 65, and the inherent resistance, inductance, and capacitance of thewelding equipment and the arc.

Free wheeling diode 61 also protects the weld selector 16 from theapplication of a negative voltage to studs 17 and 18. If a negativevoltage is applied to studs 17 or 18, a large current will flow throughresistor 64, inductor 62, diode 61, conductor 23, and fuse 24. Fuse 24is rated at 150 amps. The large current flow will blow fuse 24, therebyisolating and protecting the circuitry in weld selector 16.

Weld selector station 16 has three current limiting circuits: peakcurrent detector 80, overcurrent detector 73, and short circuit detector72. These current limiting circuits are explained in detail below. Thepeak current detector 80 causes switching, shutdown and driver circuit37 to momentarily turn off transistors 56a-56f if the instantaneouscurrent exceeds 2000 amps.

Overcurrent detector 73 causes driver circuit 37 to turn off transistors56a-56f if the average current exceeds 300 amps for more than onesecond. Short circuit detector 72 causes driver circuit 37 to turn offtransistors 56a-56f if the average voltage on studs 17 or 18 drops belowapproximately 4 volts. It will be appreciated that overcurrent detector73 and short circuit detector 72 interact. Assume that a low valueresistance is placed across either stud 17 or 18 and stud 19 so that anaverage current greater than 300 amps flows out of stud 17 or 18. Theovercurrent detector 73 will cause transistors 56a-56f to be turned off.The current will then drop and, the voltage between stud 17 or 18 andstud 19 will drop to a low value. Short circuit detector 72 will thenkeep transistors 56a-56f in the off state until the low value resistanceis removed.

The circuit power supplies 38 provides ±14 VDC, ±7 VDC, and +15 VDC,each with an isolated return, and also provides +66.7 VDC and +1.0 VDC,both having conductor 23 as a common return. Power supplies 38 isdescribed in detail below.

Switching, shutdown and driver circuit 37 is controlled by currentdetectors 72, 73 and 80, pulsewidth modulators 32 and 34, and STICK &TIG control circuit 30. Driver circuit 37 turns transistors 56a-56f onand off, as required. The operation of driver circuit 37 is described indetail below.

It will be appreciated that different types of welding require differentvoltage/current characteristics. The STICK & TIG mode of welding is bestperformed with a constant current source.

STICK & TIG control circuit 30 senses and compensates for the currentflowing through resistance bar 25 so that, if the STICK & TIG mode isselected, weld selector station 16 will approximate a constant currentsource.

If the STICK & TIG control circuit 30 is not selected, weld selector 16and stud 18 approximate a constant voltage source, which is preferredfor MIG & PULSE type welding.

As previously described, weld selector station 16 is a switching powersupply, with a selectable switching frequency of 10 to 1500 Hz. It willbe appreciated that the type of welding and the welding gas andmaterials used require different switching frequencies, waveforms, andvoltage-current characteristics for best performance. Pulsewidthmodulators 32 and 34 allow the welder to select these parameters. Bothmodulators 32 and 34, contain circuitry which increases the outputpulsewidth if the input voltage, VIN, on conductor 14 decreases, so thatthe energy supplied to the weld remains approximately constant over alarge (30 volts to 150 volts) range of input voltage VIN.

Low frequency modulator 32 provides an output signal with a range of 10to 400 Hz. High frequency modulator 34 provides an output signal with arange of 400 to 1500 Hz. The outputs of modulators 32 and 34 can also becombined to provide a low frequency (5 to 400 Hz) output which ischopped at a high frequency (400 to 1500 Hz). This chopping actionprevents inductors 62 and 70 from saturating, thereby improving theoutput regulation provided by them. The output of either modulator 32 or34 can be combined with the output of STICK & TIG control circuit 30 toprovide a pulsed constant current source. In the preferred embodiment,only one or two of the circuits 30, 32 and 34 can be activated at anyone time.

The feeder control circuit 99 controls the speed at which the weldingmaterial is fed (inched) into the weld. Two commonly used feeder controlcircuits are the arc voltage feeder, which typically requiresapproximately 20 to 40 volts, and the DC controlled feeder, whichtypically requires approximately 30 to 50 volts. High/low voltageswitchover circuit 86 automatically selects the voltage on one ofconductors 71, 82, 91 and 94 to maintain the proper operating voltagefor feeder control circuit 99.

A typical feeder control circuit 99 will also have a potentiometer forvarying the rate at which the welding material is inched into the weld.However, it will be appreciated that the welder will desire a much lowerinch feed speed when he is adjusting the extended length of the weldingmaterial than the inch feed speed desired during welding. High/lowvoltage switchover circuit 86 automatically switches the inch feed speedfrom the higher rate during welding to a lower rate when welding is notactually being performed.

Turn now to FIG. 3, which is a schematic diagram of the switching,shutdown and driver circuit 37. Driver circuit 37 receives +7 volts onconductor 100 and -7 volts on conductor 102, both referenced to a 7 vreturn conductor 101, and receives +14 volts on conductor 104 and -14volts on conductor 106, both referenced to a 14 v return conductor 105.

A first positive input of driver 37 is connected to conductor 35, andthe first negative input to a first signal return conductor 36. Thesecond positive input of driver 37 is connected to conductor 81, and thesecond negative input to a second signal return conductor 75. The outputof driver 37 is connected to conductor 40, referenced to output returnconductor 41. Driver 37 may be thought of, in a grossly over-simplifiedmanner, as an electrically isolated two-input AND gate with the secondinput, conductor 81, being inverted. Therefore, if a logic 0 voltage ispresent on conductor 35 and/or a logic 1 is present on conductor 81, theoutput voltage on conductor 40 will be a logic 0, selectable as 0 voltsor -6 volts. Conversely, if a logic 1 voltage is present on conductor 35and a logic 0 is also present on conductor 81, the output voltage onconductor 40 will be a logic 1, approximately +6 volts.

Referring briefly to FIG. 2, it will be seen that conductor 40 isconnected to the bases of transistors 56a-56f, and conductor 41 isconnected, through resistors 57a-57f, to the emitters of transistors56a-56f. Therefore, a logic 1 (+6 volts) on conductor 40 will turn ontransistors 56a-56f, and a logic 0 will turn off transistors 56a-56f. Itwill be appreciated that the stored charge in the bases of transistors56a-56f will have a substantial effect on their turnoff time. Some typesof welding require that the turnoff be rapid, whereas other types ofwelding require a longer turnoff time. As will be shown below, driver 37has a selectable logic 0 output of 0 volts, which provides a longerturnoff time, or -6 volts, which rapidly sweeps away the base charge andprovides a rapid turnoff time.

Going back to FIG. 3, input signal conductor 81 is connected throughcurrent limiting resistor 110 and conductor 111 to the anode (PIN 1) ofthe LED of isolator 112. Signal return conductor 75 is connected to thecathode (PIN 2) of the LED of isolator 112. Isolator 112 is aschmitt-trigger, inverting, open collector optoisolator such as theMOC-5008, manufactured by Motorola, Inc., Phoenix, Ariz. Isolator 112contains a light emitting diode (LED), a photodetector, aschmitt-trigger, and an open collector output transistor. The groundterminal (PIN 5) of isolator 112 is connected to the negative supplyconductor 114. The VCC terminal (PIN 6) of isolator 112 is connected tothe output of a regulator 130 and the positive terminal of a filtercapacitor 132, by conductor 113. The output of regulator 130 isnominally +5 volts with respect to conductor 114 and provides the 5 voltpower required for operation of isolator 112. The negative input ofregulator 130 and the negative terminal of filter capacitor 132 areconnected to the negative supply conductor 114. The positive input ofregulator 130 is connected to the 14 V return conductor 105. Theoperation of regulator 130 is well known to those skilled in the art.

The output (PIN 4) of isolator 112 is connected through the seriescombination of conductor 115, blocking diode 116, and conductor 117 tothe cathode of a 3.6 volt zener diode 120, one end of a resistor 121,and to the enable input (PIN 2) of a half-bridge driver 133. A logic 1on conductor 81 causes isolator 112 to place a logic 0 on conductor 117.

The anode of zener diode 120 is connected to the negative supplyconductor 114. The purpose of zener diode 120 is to assure that thevoltage on conductor 117 does not exceed the maximum rated input voltageof half-bridge driver 133.

In the inventor's prototype of the present invention, driver circuit 37was in close proximity to the power supply circuits 38. It wasdetermined that transistors from the power supply were being coupledinto isolator 112, thereby affecting its function. Diode 116 serves toprevent these transients from affecting isolator 112. Diode 116 is aSchottky diode so that the logic 0 voltage provided by isolator 112 onconductor 117 will be within the input voltage specification forhalf-bridge driver 133.

Conductor 35 is connected to one end of a loading resistor 122 and tothe anode (PIN 1) of the LED of isolator 123. Signal return conductor 36is connected to the other end of resistor 122 and to the cathode (PIN 2)of the LED of isolator 123. Isolator 123 is an optoisolator with twoout-of-phase output transistors such as the M57215L, manufactured byMitsubishi Electric Company, Tokyo, Japan. The ground terminal (PIN 4)of isolator 123 is connected to the negative supply conductor 114. Thefirst VCC terminal (PIN 3) is connected to the +5 volt output ofregulator 130 by conductor 113. The second VCC terminal (PIN 8) isconnected to the 14 v return conductor 105.

The collector of the first output transistor (PIN 7) is connected to the14 v return conductor 105 through current limiting resistor 124. Theemitter of the first output transistor (PIN 6) and the collector of thesecond output transistor (PIN 5) are connected together, to the otherend of resistor 121, and to one end of pulldown resistor 127 byconductor 126. The other end of resistor 127 is connected to thenegative supply conductor 114. A logic 0 on conductor 35 turns off thefirst output transistor and turns on the second output transistor ofisolator 123, thereby placing a logic 0 on conductor 126.

Since isolator 112 has an open collector output it can only place eithera logic 0 or an open circuit onto conductor 117. However, isolator 123can place a logic 0 or a logic 1 onto conductor 126. It will thereforebe appreciated that if the voltage on conductor 35 corresponds to alogic 0 and/or the voltage on conductor 81 corresponds to a logic 1 thevoltage on conductor 117 will correspond to a logic 0. Likewise, if thevoltage on conductor 35 corresponds to a logic 1 and the voltage onconductor 81 corresponds to a logic 0, the voltage on conductor 117 willcorrespond to a logic 1.

As previously stated, conductor 117 is connected to the enable input(PIN 2) of a half-bridge driver 133 such as the SG3635A manufactured bySilicon General, Garden Grove, Calif. The pulse input (PIN 5) and theground terminal (PIN 3) of driver 133 are connected to negative supplyconductor 114. The VCC input (PIN 1) of driver 133 is connected topositive supply conductor 136. Driver 133, as configured, functions as avoltage-shifting inverter.

The output (PIN 4) of driver 133 is connected by conductor 134 to oneend of a 3.3 kilohm load resistor 135, the cathode of 15 volt zenerdiode 145, the cathode of 20 volt zener diode 163, the anode of 20 voltzener diode 152, the source of transistors 149 and 160, one end of a 25ohm load resistor 164, and output signal conductor 40. The 14 v returnconductor 105 is connected to the other end of resistor 135, to one endof switch 153, to the cathode of 15 volt zener diode 144 and to thewiper and one end of 400 ohm potentiometer 146. The anode of zener diode144 is connected to the anode of zener diode 145. Zener diodes 144 and145 assure that the voltage on conductor 134 does not exceed thegate-to-source voltage of transistors 149 and 160. The other end ofpotentiometer 146 is connected to the gate of transistor 149 byconductor 148 through a damping ferrite bead 147. The other end ofswitch 153 is connected by conductor 154 to the wiper and one end of 400ohm potentiometer 155. The other end of potentiometer 155 is connectedto the gate of transistor 160 by conductor 157 through damping ferritebead 156.

The drain of transistor 149 is connected to one end of a 0.425 ohmpotentiometer 151 by conductor 150. The wiper and other end ofpotentiometer 151 and the cathode of zener diode 152 are connected tothe +7 volt conductor 100. The drain of transistor 160 is connected toone end of a 0.05 ohm potentiometer 162 by conductor 161. The wiper andother end of potentiometer 162 and the cathode of zener diode 163 areconnected to the -7 volt conductor 102. The 7 v return conductor 101 isconnected to the other end of resistor 164 and the output signal returnconductor 41. Diodes 152 and 163 limit the drain-to-source voltage ontransistors 149 and 160, respectively.

Transistor 149 is an N-channel enhancement mode power MOSFET such as theIRF250 manufactured by International Rectifier Corporation, Los Angeles,Calif. Transistor 160 is a P-channel enhancement mode power MOSFET suchas the IRF9130, also manufactured by International RectifierCorporation. As configured, transistors 149 and 160 function as a powerinverter.

Assume that the voltage at the enable input of half-bridge driver 133corresponds to a logic 1; driver 133 will place approximately -14 voltson conductor 134 with respect to conductor 105. This turns on transistor149 and turns off transistor 160, thereby placing approximately +6 voltson conductor 41 with respect to conductor 41. It will be appreciatedthat this turns on transistors 56a-56f of FIG. 2.

Assume now that the voltage at the enable input of driver 133corresponds to a logic 0; driver 133 will place approximately +14 voltsonto conductor 134 with respect to conductor 105. Assume also thatswitch 153 is closed. This turns off transistor 149 and turns ontransistor 160, thereby placing approximately -6 volts on conductor 40with respect to conductor 41. It will be appreciated that this rapidlyturns off transistors 56a-56f of FIG. 2 by rapidly sweeping away anystored base charge. However, if switch 153 is open, then bothtransistors 149 and 160 will be turned off, thereby placing 0 volts onconductor 40 with respect to conductor 41. It will be appreciated thatthis will turn off transistors 56a-56f of FIG. 2, but at a slower ratesince the stored base charge decays at a slower rate.

The negative supply conductor 114 is connected to the negative end ofcapacitor 141 and to one end of potentiometer 143. The wiper and theother end of potentiometer 143 are connected to the -14 volt supplyconductor 106. The positive supply conductor 136 is connected to thepositive end of capacitor 140 and to one end of potentiometer 142. Thewiper and other end of potentiometer 142 are connected to the +14 voltsupply conductor 104. The other ends of capacitors 140 and 141 areconnected to the 14 v return conductor 105.

It will be appreciated that potentiometers 142 and 143 vary the maximumpositive and negative voltage, respectively, that driver 133 places onconductor 134 and therefore vary the on-resistance of transistors 160and 149, respectively.

It will be noted that transistors 160 and 149 have a gate-to-sourceinput capacitance of approximately 2000 picofarads. It will beappreciated that potentiometers 155 and 146 vary the rate at which thiscapacitance can be charged or discharged and therefore vary the rate attransistors 160 and 149, respectively, turn on and off.

Potentiometers 151 and 162, which are in series with the drain oftransistors 149 and 160, respectively, vary the maximum currentavailable to turn transistors 56a-56f of FIG. 2 on and off,respectively.

It will therefore be appreciated that potentiometers 142, 143, 146, 155,151 and 162, and switch 153 allow the user to vary the switchingcharacteristics of transistors 56a-56f to obtain the arc characteristicsdesired for the type of welding being done.

Returning to FIG. 2, switching, shutdown and driver circuit 37 istherefore controlled by the outputs on current detectors 72, 73 and 80on conductors 81 and 75, and by the output of pulse modulator 34 onconductors 35 and 36.

Resistor 164 provides additional loading for transistors 149 and 160 andcircuit power supplies 38 so that the output voltage on conductor 40,with respect to conductor 41, does not exceed the allowable base-emittervoltage of transistors 56a-56f of FIG. 2.

Turn now to FIG. 4, which is a schematic diagram of peak currentdetector 80, overcurrent detector 73, and short circuit detector 72.Referring briefly to FIG. 2, peak current detector 80 causes drivercircuit 37 to turn off transistors 56a-56f whenever the current throughresistance bar 25 exceeds the maximum rated peak current for paralleledtransistor 56a-56f. In the preferred embodiment, the allowable peakcurrent is 2000 amps. Regulator 170 steps the +15 volts present onconductor 440 down to +5 volts. The ground of regulator 170 is connectedto the 15 v return conductor 441. The output of regulator 170 isconnected to a filter capacitor 172, the VCC terminal (PIN 8) of currentsense latch 176 and the driver circuit input (FIG. 2) by conductor 81.

Current sense latch 176 is a latching comparator with a low, 0.1 volt,threshold, such as the SG1549 manufactured by Silicon General, GardenGrove, Calif. The "HI CM" inputs (PINS 1 and 2) are not used and areconnected to each other by conductor 177. Current signal input conductor76 is connected to one end of 47 ohm resistor 173, one end of capacitor174, and to the "LO CM +" input (PIN 3) of latch 176. The current signalreturn conductor 77 is connected to one end of a 500 ohm potentiometer175. The wiper and the other end of potentiometer 175 are connected bythe 15 v return conductor 441 to the other end of resistor 173, theother end of 0.02 microfarad capacitor 174, and the "LO CM -"/ground(PIN 4) of latch 176. Resistor 173 and potentiometer 175 form a variablevoltage divider for calibrating the peak current trip level. Capacitor174 filters out high frequency noise.

The open collector output (PIN 5) of latch 176 is connected to conductor75. Therefore, when the input voltage (PIN 3) exceeds approximately 0.1volt with respect to conductor 441, conductor 75 will be shorted toconductor 441. Since conductor 81 is connected to the +5 volt output ofregulator 170, conductor 81 will be at +5 volts (a logic 1) with respectto conductor 75. As previously discussed, a logic 1 on conductor 81 willcause driver 37 to turn off transistors 56a-56f (FIG. 2).

Referring again to FIG. 2, the current through transistors 56a-56f alsoflows through resistance bar 25 and develops a voltage drop acrossresistance bar 25. Conductors 76 and 77 present a portion of thisvoltage to latch 176. The value of resistance bar 25 between conductors76 and 77 is approximately 0.1 volt/2000 amps=50 microohms. When 2000amps flows through transistors 56a-56f, the voltage between conductors76 and 77 will be approximately 0.1 volts, thus setting latch 176 (FIG.4) and turning off transistors 56a-56f. It will be appreciated that alatching device was chosen for device 176 so that a well-definedshutdown of transistors 56a-56f occurs when the peak current rating isexceeded. If a latch was not used for device 176 then, when the peakcurrent exceeded the safe value, transistors 56a-56f would be turnedoff; this would cause the current to quickly drop below the safe value,which would cause transistors 56a-56f to turn back on again, causing thecurrent to quickly rise above the safe value. The net result would be ahigh frequency oscillation with an average current, instead of a peakcurrent, of 2000 amps.

Turning back to FIG. 4, since device 176 is a latch there must be ameans to reset it. Although a manual reset could be used, an automaticreset is preferable. The output of pulse clock 181 is connected byconductor 180 to the reset input (PIN 7) of latch 176. In the preferredembodiment of the present invention pulse clock 181 has a frequency of60 Hz. The frequency of pulse clock 181 is not critical but should befast enough to allow welding to continue without a noticeable delay. Theperiod of the pulse is not critical, but must be at least 150microseconds for latch 176 specified to be reset.

Referring briefly to FIG. 2, short circuit detector 72 monitors thevoltage on STICK & TIG stud 17 and, through resistor 66 and inductor 70,the voltage on MIG & PULSE stud 18. If the voltage on stud 17 or 18drops below a predetermined trip voltage because of a short circuit orbecause overcurrent detector 73 has tripped, short circuit detector 72causes switching, shutdown and driver circuit 37 to turn off transistors56a-56f. Short circuit detector 72 will then keep transistors 56a-56foff until the short is removed and the voltage at studs 17 and 18 risesabove the trip voltage.

Since transistors 56a-56f are now off, they cannot raise the voltage onstuds 17 and 18, even after the short is removed. The restart voltage isprovided by the 2.4 ohm arc sustaining resistor 45. It will beappreciated that resistor 45 cannot pass sufficient current to raise thevoltage at studs 17 and 18 until the short circuit is removed. Once theshort circuit is removed, resistor 45 will pull studs 17 and 18 abovethe trip voltage and short circuit detector 37 will allow driver circuit37 to turn on transistors 56a-56f so that welding can start again.

Although the input of short circuit detector 72 is shown connected tothe STICK & TIG stud 17, it will be appreciated that the input may beconnected to any point in the circuit that will have a low voltage whenstud 17 or 18 is shorted. Two of the most likely other points are MIG &PULSE stud 18, and conductor 67 between resistor 66 and inductor 70.

Turning back to FIG. 4, conductor 71 is connected to the anode ofblocking diode 191. The cathode of blocking diode 101 is connected byconductor 192 to one end of 500 ohm resistor 193 and to the positiveterminal of 680 microfarad capacitor 194. Diode 191 allows capacitor 194to charge, but not to discharge, through conductor 71. The other end ofresistor 193 is connected by conductor 196 to one end of the 12 volt,400 ohm coil of relay 197 and to the cathode of 15 volt zener diode 195.The other end of the coil of relay 197, the anode of zener diode 195,and the negative terminal of capacitor 194 are connected to the 80 vreturn conductor 23.

Since the coil of relay 197 is rated at 12 volts and the voltage onconductor 71 can be 80 volts or more, resistor 193 and zener diode 195serve to limit the voltage on the coil of relay 197.

During normal welding the voltage on conductor 71 will be sufficient tokeep relay 197 energized. However, if stud 17 and/or stud 18 are shortedto stud 19, the voltage on conductor 71 will be inadequate to keep relay197 energized. Capacitor 194 serves as a holding capacitor to keep relay197 energized for a short period of time. This prevents short circuitdetector 72 from detecting brief shorts. During a brief short, peakcurrent detector 80 provides protection.

If a prolonged short occurs, capacitor 174 will discharge throughresistor 193, diode 195, and the coil of relay 197, and relay 197 willbe de-energized.

Relay 197 has a pair of normally closed contacts. If a short is notpresent, the voltage on conductor 71 will be adequate to keep relay 197energized, and the contacts will be open. If a prolonged short occurs,relay 197 will drop out and the contacts of relay 197 will close. Onecontact is connected to 15 v conductor 441 and the other contact isconnected to conductor 75.

When a prolonged short occurs, relay 197 will drop out, the contactswill close, and conductor 75 will be connected to conductor 441.Conductor 81 will therefore be at +5 volts (a logic 1) with respect toconductor 75. As previously described, a logic 1 on conductor 81 causesdriver circuit 37 (FIG. 2) to turn off transistors 56a-56f (FIG. 2).

This condition will remain until, as described above, the short isremoved and the voltage on studs 17 and 18 rises above the trip voltage.

Returning briefly to FIG. 2, overcurrent detector 73 monitors thecurrent flowing through transistors 56a-56f and turns them off when theaverage current exceeds the maximum rated average current of transistors56a-56f. In the preferred embodiment of the present invention,overcurrent detector 73 is set to trip at 250 to 300 amps. Assuming thatthe current flow is 300 amps, then there will be approximately 0.03ohms×300 amps=9 volts developed across resistor 64. This voltage ispresented to overcurrent detector 73 by conductors 63 and 65.

Turning now to FIG. 4, conductor 63 is connected to one end of 25 ohmpotentiometer 182. The wiper of potentiometer 182 is connected byconductor 183 to the positive end of 4700 microfarad capacitor 186 andto one end of the 400 ohm coil of relay 190. Conductor 65 is connectedto the other end of potentiometer 182 and, through the seriescombination of diode 184, 47 ohm resistor 185, and conductor 187, to theother end of capacitor 186 and the other end of the coil of relay 198.

The voltage placed between conductors 63 and 65 will, when the currentexceeds the trip value, cause relay 190 to pull in. Potentiometer 182allows the trip point to be set to match the average current ratings oftransistor 56a-56f (FIG. 2). Resistor 185 is a current limiting resistorand diode 184 prevents capacitor 186 from discharging throughpotentiometer 182 and resistor 64.

One contact of a pair of normally open contacts in relay 190 isconnected to conductor 75. The other contact in relay 190 is connectedto the 15 v return conductor 441. When the current flowing throughtransistors 56a-56f (FIG. 2) is less than the trip value relay 190 willnot be energized and the contacts will be open. However, when thecurrent exceeds the trip value, relay 190 will be energized and thecontacts of relay 190 will be closed, thereby connecting the 15 v returnconductor 441 to conductor 75. This places +5 volts, a logic 1, onconductor 81 with respect to conductor 75 and, as previously described,a logic 1 on conductor 81 causes driver circuit 37 (FIG. 2) to turn offtransistors 56a-56f (FIG. 2). The current then falls below the tripvalue. Capacitor 186 and the resistance of the coil of relay 190 have atime constant of approximately two seconds. This assures thattransistors 56a-56f (FIG. 2) are off for a sufficient time to cause theoutput current and voltage to fall to zero.

Returning to FIG. 2, it will be appreciated that if overcurrent detect73 is tripped, the current flow will be interrupted for around twoseconds. This time period is more than adequate for the voltage on studs17 and 18 to fall below the trip voltage of short circuit detector 72.Therefore, even after the current falls and overcurrent detector 73 hasreset, short circuit detector 72 will keep transistors 56a-56f turnedoff until the short is removed and resistor 45 pulls the voltage onstuds 17 and 18 back above the trip voltage.

Continuing with FIG. 2, STICK/TIG control circuit 30 monitors thecurrent out of studs 17 and 18 and causes weld selector 16 to functionas a constant current source. Current flowing out of studs 17 and 18must flow through resistance bar 25. One of the taps of resistance bar25 is connected to positive signal conductor 26. Another of the taps ofresistance bar 25 is connected to the negative signal conductor 27.Conductors 26 and 27 are connected to the inputs of control circuit 30.

The voltage output of control circuit 30 on conductor 31 increases asthe current through resistance bar 25 decreases. This causes pulsewidthmodulator 34 to increase the on-time of its output on conductors 35 and36. This causes driver circuit 30 to increase the on-time of its outputon conductors 40 and 41, thereby increasing the on-time of transistors56a-56f, which increase the output current.

Turn now to FIG. 5, which is a schematic diagram of STICK/TIG controlcircuit 30. Control circuit 30 is powered by the +13.3 volt differencebetween +80 volt conductor 21 and +66.7 volt conductor 430, whichfunctions as the circuit ground for control circuit 30. The positivesignal conductor 26 is connected to one end of 100 ohm resistor 221. Theother end of resistor 221 is connected by conductor 222 to one end of100 ohm resistor 212, the positive terminal of 100 microfarad capacitor213, and to one end of 30 kilohm resistor 223. The other end of resistor223 is connected to one end of 2 kilohm potentiometer 224. The other endof potentiometer 224 is connected through 50 kilohm resistor 225 tocircuit ground 430. The wiper of potentiometer 224 is connected byconductor 226 to the non-inverting input of operational amplifier(op-amp) 217.

The negative signal conductor 27 is connected to one end of 100 ohmresistor 210. The other end of resistor 210 is connected by conductor211 to the other end of resistor 212, the other end of capacitor 213,and to one end of 50 kilohm resistor 214. The other end of resistor 214is connected by conductor 215 to one end of 500 kilohm feedback resistor216, the inverting input of op-amp 217, and to one end of 50 kilohmresistor 220. The other end of resistor 216 is connected to the output(PIN 1) of op-amp 217 by conductor 227. The other end of resistor 220 isconnected to circuit ground 430.

It will be appreciated that op-amp 217 is configured as adifferential-input, negative-feedback gain stage with a gain controlpotentiometer 224. Capacitor 213 is a smoothing capacitor whicheliminates high frequency transients.

The output of op-amp 217 is also connected by conductor 227 to one endof 5.1 kilohm stabilizing resistor 232, and one end of 5.1 kilohmresistor 240. The other end of resistor 240 is connected to circuitground 430 through 5.1 kilohm resistor 241, and is connected byconductor 242 to the inverting input of op-amp 246 and one end of 30kilohm potentiometer 243.

The ground input of op-amps 217 and 246 are connected to circuit ground430. Conductor 21 is connected to the VCC input of op-amps 217 and 246,and one end of 200 ohm resistor 230. The other end of resistor 230 isconnected by conductor 231 to the other end of resistor 232, to theanode of temperature compensating diode 233, and one end of 5.1 kilohmresistor 235. The cathode of diode 233 is connected to the cathode of6.8 volt zener diode 234. The anode of zener diode 234 is connected tocircuit ground 430. The other end of resistor 235 is connected to thenon-inverting input of op-amp 246 by conductor 237, and connected tocircuit ground 430 through 5.1 kilohm resistor 236.

Because of the temperatures encountered in weld selector 16, resistors214, 216, 220, 223, 225, 232, 235, 236, 240 and 241 should be metal filmresistors so that the output of STICK/TIG control circuit 30 will not beoverly temperature sensitive.

It will be appreciated that diodes 233 and 234 form atemperature-compensated 7.5 volt reference and that resistors 235 and236 place one-half of this reference voltage onto the non-invertinginput of op-amp 246.

The output of op-amp 246 is connected by conductor 247 to one end andthe wiper of 6.8 kilohm current gain limit potentiometer 244, to circuitground 430 through 6.8 kilohm load resistor 245, to one end of currentlimiting 10 ohm resistor 250, and to one end of 100 kilohm resistor 252.The other end of limit potentiometer 244 is connected to the wiper andthe other end of current gain potentiometer 243. The other end ofresistor 250 is connected to circuit ground 430 through 1000 microfaradcapacitor 251. The other end of resistor 252 is connected to the anodeof blocking diode 253. The cathode of diode 253 is connected to outputconductor 31 through switch 254.

Op-amp 246 is configured as a variable gain amplifier with a referencevoltage (3.75 volts) connected to the non-inverting input and the outputof op-amp 217 connected to the inverting input. The output of op-amp 246on conductor 247 is therefore proportional to the negative of thedifferential voltage between input conductors 26 and 27.

Potentiometers 243 and 244 control the gain of op-amp 246. Potentiometer243 may be set by the welder to obtain the desired arc characteristics.Potentiometer 244 is generally not accessible to the welder and is usedto set the minimum gain of amplifier 246.

Capacitor 251 and resistor 250 heavily load the output of op-amp 246 sothat the voltage on conductor 247 changes at a relatively slow rate.This assures that the output of control circuit 30 is responsive tothose variations in the welding current caused by welding conditions andnot responsive to those variations caused by the normal operation ofpulsewidth modulators 32 and 34.

Returning briefly to FIG. 2, modulator 32 is a low frequency pulsewidthmodulator. In the preferred embodiment, the pulse frequency is variablefrom approximately 10 to 400 Hz, and the duty cycle is variable from 0to over 90 percent. These values are not critical but do allow thewelder considerable latitude for obtaining the most desirable arccharacteristics.

Modulator 32 is input voltage compensated. The pulsewidth increases asthe input voltage on conductor 21 decreases so that the average powerdelivered to the weld is unaffected by input voltage variations.

Turn now to FIG. 6, which is a schematic diagram of low frequencypulsewidth modulator 32. Modulator 32 is powered by the 13.3 voltdifference between +80 volt conductor 21 and +66.7 volt conductor 430,which also serves as the circuit ground. Conductor 21 is connected to0.1 microfarad capacitor 260 and to the VCCIN terminal of modulator 261.Modulator 261 is a regulating pulsewidth modulator such as the LM3524manufactured by National Semiconductor Corporation, Santa Clara, Calif.

Modulator 261 contains a +5 volt regulator 262, an errortransconductance amplifier 264, a current limit amplifier 265, anoscillator 266, a shutdown control 267, a comparator 271, anedge-triggered toggle 272, two 3-input NOR gates 275 and 276, and twoNPN output transistors 277 and 280.

The output of transconductance amplifier 264, current limit amplifier265, and shutdown control 267 are connected to the inverting input ofcomparator 271 by conductor 270. Current limit amplifier 265 andshutdown control 267 are each able to override the output oftransconductance amplifier 264. Current limit amplifier 265 is not usedso its inverting input and non-inverting input are connected to circuitground 430 by conductor 286 and conductor 287, respectively. Shutdowncontrol 67 is also not used so its input is connected to circuit ground430 by conductor 290.

Regulator 262 steps the +13.3 volts present on conductor 21 (withrespect to circuit ground 430) down to +5 volts and places thisregulated +5 volts onto conductor 263. This +5 volts also powers some ofthe internal circuitry of modulator 261. The input of regulator 262 isconnected to conductor 21. The ground terminal of modulator 261 isconnected by conductor 285 to circuit ground 430.

The output of oscillator 266 is connected by conductor 273 to the toggleinput of toggle 272 and one input of NOR gates 275 and 276. The Q outputof toggle 272 is connected by conductor 281 to a second input of NORgate 275. The negated Q output of toggle 272 is connected by conductor282 to a second input of NOR gate 276. The output of comparator 271 isconnected by conductor 274 to the third input of NOR gates 275 and 276.The outputs of NOR gates 275 and 276 are connected to the bases oftransistors 277 and 280, respectively. The collectors of transistors 277and 280 are connected to +5 volt conductor 263. The emitters oftransistors 277 and 280 are connected to output signal conductor 33.

The R input of oscillator 266 is connected to circuit ground 430 throughthe series combination of conductor 297 and 500 kilohm potentiometer301. The C input of oscillator 266 is connected by conductor 296 to oneend of 0.1 microfarad capacitor 300, and the non-inverting input ofcomparator 271. The other end of capacitor 300 is connected to circuitground 430. Capacitor 300 and potentiometer 301 determines the frequencyof oscillator 266.

The output of oscillator 266 on conductor 273 is a pulsed waveform 269which, during the pulse period, toggles the toggle 272 and, through NORgates 275 and 276, turn off transistors 277 and 280, respectively.Oscillator 266 also causes a sawtooth voltage 268 to be present onconductor 296. It will be appreciated that when the voltage on conductor296 rises above the voltage on conductor 270, the output of comparator271 will be a logic 1, thereby turning off transistors 277 and 280through NOR gates 275 and 276, respectively.

The regulated +5 volt conductor 263 is connected to one end of 0.1microfarad filter capacitor 291 and to one end of 5.1 kilohm resistor292. The other end of capacitors 260 and 291 are connected to circuitground 430. The other end of resistor 292 is connected by conductor 283to the inverting input of transconductance amplifier 264 and one end of10 ohm resistor 293. The other end of resistor 293 is connected byconductor 284 to the non-inverting input of transconductance amplifier264, one end of 68 kilohm resistor 294, and one end of 178 kilohmresistor 295. The other end of resistor 294 is connected to circuitground 430. The other end of resistor 295 is connected to +1 voltconductor 431.

It will be appreciated that resistors 292, 293, 294 and 295 form avoltage divider string which provides the differential voltage input totransconductance amplifier 264. As will be described below, +66.7 voltconductor 430 is regulated at -13.3 volts with respect to +80 voltconductor 21. Neither +80 volt conductor 21 nor +66.7 volt conductor 430are regulated with respect to +1 volt conductor 431. It will also beappreciated that conductor 283 will always be positive with respect toconductor 284, therefore transconductance amplifier will always betrying to sink, rather than source, current from conductor 270.

Transconductance amplifier 264 has a maximum output of a few hundredmicroamps. As previously stated, the output of transconductanceamplifier 264 is also placed onto conductor 270. Since transconductanceamplifier 264 is connected in an open-loop configuration, and since theinverting input is always positive with respect to the non-invertinginput, it may be expected that amplifier 264 would drive conductor 270down to circuit ground 430. However, the input to amplifier 264 is thesmall voltage (millivolts) developed across 10 ohm resistor 293. Thissmall input voltage, the small output current of amplifier 264, and the100 kilohm load resistor 315 therefore force amplifier 264 to operate asa linear transconductance amplifier, even when connected in an open loopconfiguration.

It will be appreciated that the differential voltage across resistor 293will increase as the voltage on conductors 21 and 430 increases withrespect to +1 volt conductor 431. This causes amplifier 264 to sink morecurrent and pulls the voltage on conductor 261 down so as to decreasethe width of the output pulses. It will be appreciated that this, inturn, causes the average current through transistors 56a-56f (FIG. 2) todecrease, thereby compensating for the increase in the input voltage on+80 volt conductor 21.

The output of transconductance amplifier 264 is also connected byconductor 270 to one end of 0.1 microfarad capacitor 316 and one end of100 kilohm resistor 315. The other end of resistor 315 is connected tothe cathode of blocking diode 314. The anode of diode 314 is connectedby conductor 311 to one end of 12 kilohm resistor 313, one end of 1.0microfarad capacitor 312, and to one end of switch 310. The other end ofcapacitors 312 and 316 and the other end of resistor 313 are connectedto circuit ground 430.

The other end of switch 310 is connected by conductor 307 to one end ofparalleled potentiometers 305 (250 kilohms) and 306 (100 kilohms). Thewipers and other ends of potentiometers 305 and 306 are connected byconductor 304 to one end of 25 kilohm potentiometer 303. The wiper andother end of potentiometer 303 are connected to +80 volt conductor 21through 22 kilohm resistor 302. It will be appreciated that, when switch310 is closed, resistors 302 and 313 and potentiometers 303, 305, and306 form an adjustable voltage divider with filtering capacitors 312 and316 and an output on conductor 270.

Conductor 270 is connected to the inverting input of comparator 271 andsets a reference level. The ramp voltage 268 on conductor 296 startsbelow the reference level on conductor 270 and so the output ofcomparator 271 is a logic 0, thus turning on either transistor 277 or280, depending upon the state of the Q and negated Q outputs of toggle272. After the voltage on conductor 296 rises above the reference levelon conductor 270, the output of comparator 271 is a logic 1, thusturning off both transistors 277 and 280. The voltage on conductor 296continues rising until it reaches the internal reference level ofoscillator 266, at which point oscillator 266 discharges capacitor 296thus causing the voltage on conductor 296 to fall to zero, and alsoplaces a pulse on conductor 273 which toggles the toggle 272 and alsokeeps transistors 277 and 280 turned off. The voltage on conductor 296begins rising again and the cycle repeats.

It will now be appreciated that the voltage on conductor 270 sets thewidth of the output pulses on conductor 33. Potentiometers 303 and 305are used to set the maximum and minimum pulsewidth points. Potentiometer306 allows the welder to vary the pulsewidth between the maximum andminimum points to obtain the desired arc characteristics.

If switch 30 is open, conductor 270 will be pulled to circuit ground 430by amplifier 264, and the output pulsewidth of modulator 261 will dropto zero. Switch 310 therefore serves as the ON/OFF switch for lowfrequency pulsewidth modulator 32.

Turn now to FIG. 7, which is a schematic diagram of high-frequencypulsewidth modulator 34. With a few exceptions, described below, theoperation of high frequency pulsewidth modulator is the same as theoperation of low frequency pulsewidth modulator 32 and components 320through 376 are identical to components 260-316, respectively, of FIG.6.

In the preferred embodiment the pulse frequency is variable from 400 to1500 Hz, the duty cycle is variable from 0 to over 90 percent, and pulsefrequency potentiometer 361 has a value of 50 kilohms. The frequencyrange of high frequency pulsewidth modulator 34 is not critical and wasselected to provide for anticipated welding needs. However, weldselector station 16 has been successfully tested with modulator 34frequencies above 50 kHz. One benefit of operation at frequenciessubstantially in excess of 1500 Hz is that the inductance, and thereforethe size, of inductors 62 and 70 may be reduced without adverselyaffecting the regulating ability of weld selector station 16.

Five volt conductor 323 is connected to circuit ground 430 through 5kilohm potentiometer 384. The wiper of potentiometer 384 is connected tothe cathode of blocking diode 383. The anode of blocking diode 383 isconnected to conductor 330. This sets an upper limit on the outputpulsewidth and, therefore sets a minimum time during each pulse that theoutput of modulator 321 is a logic 0. This minimum time is often calledthe deadband time.

It will be appreciated that this deadband time only affects the maximumpossible pulsewidth of the pulses generated by modulator 321 and has noeffect upon the maximum pulsewidth of the pulses generated by modulator261 (FIG. 6) or upon the period between high frequency and low frequencypulses.

The maximum pulsewidth set by potentiometers 363, 365 and 366 istypically below the maximum pulsewidth set by potentiometer 384.Although potentiometer 384 can be used to override the maximumpulsewidth time set by potentiometers 363, 365 and 366, its primarypurpose, as described below, is to override the maximum pulsewidth setby the output of STICK/TIG control circuit 30 and/or of low frequencypulsewidth modulator 32. It will be appreciated that the output of highfrequency pulsewidth modulator 34 is affected by potentiometers 363, 365and 366 only if switch 370 is closed.

It will be recalled that the output of STICK/TIG control circuit 30 is aDC voltage responsive to the current through resistance bar 25 (FIG. 2),can pull conductor 31 up, but not down (FIG. 5), and can be placed on(removed from) conductor 31 by closing (opening) switch 254 (FIG. 5).The output of STICK/TIG control circuit 30 is connected by conductor 31to conductor 330 of high frequency pulsewidth modulator 34.

Assume that switch 254 (FIG. 5) is closed and switch 370 is open. If theoutput current of weld selector station 16 increases, the currentthrough resistance bar 25 willincrease, and the output of STICK/TIGcontrol circuit 30 on conductor 31 will decrease. This lowers thevoltage on conductor 330, which causes modulator 321 to decrease theoutput pulsewidth, causes transistors 56a-56f to have less on time,thereby reducing the current through resistance bar 25. The result is ahigh-frequency, variable pulsewidth adjustable constant current source.

Diodes 253 (FIG. 5) and 375 form a wired-OR so, if switch 370 is closed,the output of modulator 321 will be the larger of (a) the pulsewidthdetermined by STICK/TIG control circuit 30 or (b) the pulsewidthdetermined by potentiometers 363, 365 and 366. If switch 370 is open,the output of modulator 321 will be the pulsewidth determined bySTICK/TIG control circuit 30. However, in either case, the maximumoutput pulsewidth of modulator 321 is determined, as explained above, bythe setting of deadband potentiometer 384.

The output of low frequency pulsewidth modulator 32 (FIG. 6) isconnected by conductor 33 to the pole contact of SPDT switch 380. Itwill be recalled that the output of low frequency modulator 32 is a lowfrequency pulsed output. The first position of switch 380 connectsconductor 33 to the anode of blocking diode 382 through 100 kilohmresistor 381. The cathode of diode 382 is connected to conductor 330.

Assume that switch 380 is in the first position, that STICK/TIG switch254 (FIG. 5) and HIFREQ switch 370 are open, and that LOFREQ switch 310(FIG. 6) is closed. During the period that the output of low frequencypulsewidth modulator 32 is a logic 0 the output of high frequencypulsewidth modulator 34 will also be a logic 0. During the period thatthe output of low frequency pulsewidth modulator 32 is a logic 1 theoutput of high frequency pulsewidth modulator 34 will be a highfrequency series of pulses, with the pulsewidth of each pulse beingdetermined by the setting of deadband potentiometer 384. The result isthat the output of high frequency modulator 34 will be a repeated seriesof a burst of high frequency pulses followed by a period of no outputpulses. Since both modulators 32 and 34 are input voltage compensated,weld selector station 16 (FIG. 1) approximates a high-frequency, lowburst rate, constant voltage source.

Assume now that STICK/TIG switch 254 (FIG. 5) is also closed. When theoutput of low frequency modulator 32 is a logic 0, the voltage onconductor 330, and therefore the output pulsewidth of high frequencymodulator 34, will be determined by the output of STICK/TIG controlcircuit 30, within the limit set by deadband potentiometer 384. When theoutput of low frequency modulator 32 is a logic 1, the voltage onconductor 330, and therefore the output pulsewidth of high frequencymodulator 34, will be determined by the setting of deadbandpotentiometer 384. The result is that the output of high frequencymodulator 34 will alternate between a first series of high frequencypulses whose width is determined primarily by STICK/TIG control circuit30, and a second series of high frequency pulses whose width isdetermined by deadband potentiometer 384. The duration of each serieswill be determined by low frequency modulator 32. Weld selector station16 will therefore alternate, at the low frequency rate, betweenapproximating a high pulse frequency, variable pulsewidth, constantcurrent source and a high pulse frequency, fixed pulsewidth, constantvoltage source.

Assume that switch 380 is in the second position and that switch 310(FIG. 6) is closed. Conductor 33 will be connected by switch 380 toconductor 35. It will be recalled that the output of modulator 361 andof modulator 321 can pull up, but not down. Therefore, modulators 261and 321 are configured in a wired-OR configuration. Also assume thatswitch 370 and switch 254 (FIG. 5) are open. When the output oflow-frequency modulator 32 is a logic 1, the voltage on conductor 35will be a logic 1. When the output of low frequency modulator 32 is alogic 0, the voltage on conductor 35 will also be a logic 0. The logicstate of conductor 35 will therefore follow the logic state of theoutput of low frequency modulator 32. Weld selector station willtherefore approximate an input voltage compensated, low frequency,pulsed voltage source.

Now assume that switch 370 is closed, but switch 254 (FIG. 5) is open.When the output of low frequency modulator 32 is a logic 1, the outputon conductor 35 will be a logic 1. When the output of low frequencymodulator 32 is a logic 0, the output of modulator 321 dominates, whichwill be a high frequency pulse whose pulsewidth is determined by settingof potentiometers 363, 365 and 366. The signal conductor 35 willtherefore alternate, at the low frequency rate, between a logic 1 and aburst of high frequencies pulses.

Now assume that switch 370 is open and switch 254 (FIG. 5) is closed.During the period that the output of low frequency modulator 32 is alogic 1, the output signal on conductor 35 will be a logic 1. During theperiod that the output of low frequency modulator is a logic 0, theoutput signal on conductor 35 will be a high frequency series of pulsesprovided by modulator 321. It will be recalled that, given the aboveswitch conditions, the width of the output pulses provided by modulator321 is controlled by STICK/TIG control circuit 30.

The inventor knows of no welding application wherein it would bedesirable to have switches 254 (FIG. 5), 310 (FIG. 6) and 370 (FIG. 7)all closed. Therefore, in the preferred embodiment, the operatingcondition wherein switch 254 (FIG. 5), switch 310 (FIG. 6) and switch370 (FIG. 7) are all closed is an invalid condition and turns off weldselector station 16. Methods of using mechanical and/or electricalinterlocks to prevent the simultaneous closing of all three switches254, 310, and 370 are well known to those skilled in the art and are notdetailed herein. However, some welding application unknown to theinventor may require the closing of all three switches 254, 310 and 370and therefore it may be desirable to allow this operating condition.

Returning briefly to FIG. 2, high/low voltage switchover circuit 86performs two functions: switching the speed control input of feedercontrol 99 from potentiometer 84 to a fixed resistor inside switchovercircuit 86; and regulating the power supply voltage provided to feedercontrol 99.

The welder sets potentiometer 84 to get the desired wire feed speed forthe actual welding operation. This wire feed speed is generally higherthan the wire feed speed desired when the welder is adjusting the wirefeed length prior to commencing the welding operation. Switchovercircuit 86 monitors the voltage on studs 17 and 18, which will be highwhen not welding and low when welding, and connects an internalresistor, or potentiometer 84, respectively, to feeder control 99,thereby automatically providing the more desirable wire feed speed.

A typical feeder control 99 will contain some voltage regulatingcircuitry so that minor variations in the input voltage do not affectthe wire feed speed. The input voltage for a feeder control 99 is alsotypically obtained by using a resistance voltage divider to drop the +80volts on conductor 14 down to the specified operating voltage. However,a typical resistive voltage divider may consume over 100 watts of power,which is dissipated as heat. Switchover circuit 86 eliminates thisresistive voltage divider, and therefore improves the efficiency of weldselector station 16, by switching between several voltage sourcesalready in weld selector station 16 to obtain the specified operatingvoltage.

In the preferred embodiment switchover circuit 86 can select from a tappoint on arc sustaining resistor 45, STICK & TIG stud 17, or two tappoints on load resistor 93. It will be appreciated from prior statementsthat resistor 45 already exists in weld selector station 16 to provide asustaining current for the arc when transistors 56a-56f are turned off,and that resistor 93 already exists in weld selector station 16 toprovide a minimum load for transistors 56a-56f when they are on but anarc has not been struck.

STICK & TIG stud 17 is connected to a first input of switchover circuit86 by conductor 71. Stud 17 will typically be at +80 volts before thearc is struck and +20 volts after the arc is struck. A first tap pointon resistor 93 is connected to a second input of switchover circuit 86by conductor 94. The voltage on conductor 94 will follow, but will beless than, the voltage on stud 17. A second tap point on resistor 93 isconnected to a third input of switchover circuit 86 by conductor 91. Thevoltage on conductor 91 will follow, but will be less than, the voltageon conductor 94 or stud 17. A tap point on resistor 45 is connected to afourth input of switchover circuit 86 by conductor 82. The voltage onconductor 82 will typically be +80 volts before the arc is struck and,after the arc is struck, will be between +80 volts and the voltage onstud 17.

It will be appreciated that, although stud 17 is the reference point,the voltage on stud 17 closely follows the voltage on stud 18.

Turn now to FIG. 8, which is a schematic diagram of high/low voltageswitchover circuit 86. The +80 volt conductor 21 is connected to theinput of a 15 volt regulator 390 and to one contact of a normally closedpair of contacts on relay 411. The 80 volt return conductor 23 isconnected to the ground of regulator 390, the negative terminal offilter capacitor 392, one end of filter capacitor 393, the anode of 8.2volt zener diode 395, the ground input of op-amp 397, the cathode ofblocking diode 410, one end of 0.1 microfarad smoothing capacitor 403,one end of 27 kilohm resistor 404, and one end of 250 kilohmpotentiometer 405. The regulated +15 volt output of regulator 390 isconnected by conductor 391 to the positive terminal of capacitor 392,the other end of capacitor 393, one end of 6.8 kilohm resistor 394, andto the VCC input of op-amp 397.

The other end of resistor 394 is connected by conductor 396 to thecathode of zener diode 395 and the inverting input of op-amp 397.Resistor 394 and zener diode 395 therefore form a voltage reference forop-amp 397.

The STICK/TIG stud 17 is connected by conductor 71 to the anode ofblocking diode 400. The cathode of diode 400 is connected to conductor402 through 82 kilohm resistor 401. Conductor 402 is connected to thenon-inverting input of op-amp 397, the other end of capacitor 403, theother end of resistor 404, and to the wiper and other end ofpotentiometer 405. It will be appreciated that resistors 401 and 404 andpotentiometer 405 form an adjustable voltage divider. Potentiometer 405is set so that, when the voltage on conductor 71 rises above the desiredvoltage input level of feeder control 99, the voltage on thenon-inverting input of op-amp 397 is greater than the voltage on itsinverting input. This causes the output of op-amp 397 to rise to nearthe positive rail, conductor 391. Capacitor 403 smooths out the voltageon conductor 71 so that op-amp 397 does not respond to the voltagetransients causes by the switching action of transistors 56a-56f of FIG.2.

The output of op-amp 397 is connected by conductor 406 to the cathode ofsurge protection diode 407 and to one end of the coil of relay 411. Theanode of diode 407 and the other end of the coil of relay 411 areconnected by conductor 408 to the anode of blocking diode 410. Thecathode of diode 410 is connected to the 80 volt return conductor 23.

It is well known to place a surge protection diode 407 in parallel withthe coil of a relay such as relay 411 to absorb the voltage spike whenrelay 411 is de-energized. However, even with diode 407, conductor 406may be momentarily pulled below conductor 23 by the forward voltage dropof diode 407. This momentary voltage pulse has caused erratic operationof op-amp 397. The inventor has not taken steps to ascertain whetherthis erratic operation is caused by a phantom diode or phantomtransistor in op-amp 397, or by some other characteristic. However, theinsertion of blocking diode 410 prevents this momentary voltage pulsefrom affecting op-amp 397 by isolating diode 407 and the coil of relay411 from conductor 23.

The other contact of relay 411 is connected by conductor 412 to one endof the coil of relay 420, and to one end of the coil of relay 421through 3.3 kilohm voltage reduction resistor 422. The other end of thecoil of relay 420 and relay 421 are connected to 80 volt returnconductor 23. Relays 420 and 421 each have a SPDT set of contacts.

The pole contact of relay 421 is connected by conductor 95 to the speedcontrol input of feeder control 99. The normally closed contact of relay421 is connected by conductor 423 to one end of a 12 kilohm resistor424. The other end of resistor 424 is connected to conductor 96. Thenormally open contact of relay 421 is connected by conductor 83 to oneend of 11 kilohm potentiometer 84. The wiper and other end ofpotentiometer 84 are connected by conductor 85 to conductor 96.Conductor 96 is connected to the speed control return input of feedercontrol 99.

The normally closed contact of relay 420 is connected by conductor 91 tothe second tap point on resistor 93 of FIG. 2. The normally open contactof relay 420 is connected to the pole of SP3 T switch 425. The firstposition of switch 425 connects the normally open contact of relay 420to conductor 71, which is connected to STICK/TIG stud 17 of FIG. 2. Thesecond position of switch 425 connects the normally open contact ofrelay 420 to the first tap point on resistor 93 of FIG. 2 throughconductor 94. The third position of switch 425 connects the normallyopen contact of relay 420 to the tap point on arc sustaining resistor 45of FIG. 2 through conductor 82.

The pole contact of relay 420 is connected by conductor 97 to the powervoltage input of feeder control 99. The power voltage return of feedercontrol 99 is connected by conductor 98 to the 80 volt return conductor23.

Before an arc is struck, the voltage on conductor 71 will be high,approximately 80 volts. The voltage on conductor 402 will therefore begreater than the trip voltage on conductor 396. The output of op-amp 397will then be approximately 14 volts, thereby energizing relay 411. Thisde-energizes relays 420 and 421. The second tap point on resistor 93 ofFIG. 2 will therefore be connected through conductor 91, the contacts ofrelay 420, and conductor 97 to the power voltage input of feeder control99. If feeder control 99 is specified for, say, a 40 volt input, thenthe second tap point of resistor 93 of FIG. 2 is adjusted to give 40volts on conductor 97 when the voltage on conductor 71 is 80 volts.

After an arc is struck, the voltage on conductor 71 will drop toapproximately 20 volts. The voltage on conductor 402 will fall below thetrip voltage on conductor 396 and relay 411 will be de-energized. Relays420 and 421 will be connected through conductor 412 and the contacts ofrelay 411 to +80 volt conductor 21 and therefore energized. The powervoltage input of feeder control 99 will therefore be connected throughconductor 97 and the normally open contacts of relay 420 to the pole ofswitch 425. Switch 425 is placed in a selected position so that thevoltage on conductor 97 is closest to the desired voltage, 40 volts inthis example. In the example given, the voltage on conductor 71 is only20 volts, so switch 425 positions 2 or 3 would be selected.

It will be appreciated that a different feeder control 99, such as anarc feeder control, will require a different power input voltage,typically 20 volts. In this case switch 425 would most probably be setto position 1, the voltage on conductor 71. It will also be appreciatedthat, in some applications, it may be desirable to provide anotherswitch similar to switch 425 to allow selection of different voltagepoints when relay 420 is energized. It will also be appreciated that itmay be desirable to have switch 425 connected to potentiometer 405, or aswitchable voltage divider network instead of potentiometer 405, so thata welder could select a position on switch 425 and automatically obtainthe desired feeder voltage on conductor 97 and the corresponding desiredresistance of potentiometer 425.

In a similar manner, before an arc is struck, relay 421 will bede-energized, thereby connecting resistor 424 across the speed controlinputs of feeder control 99 to yield a slow, fixed wire feed speed toallow the welder to adjust the wire length.

Likewise, after an arc is struck, relay 421 will be energized, therebyconnecting potentiometer 84 across the speed control inputs of feedercontrol 99 to yield the higher, welder-set, wire feed speed desired forthe actual welding operation. It will be appreciated that potentiometer84 may be contained within feeder control 99 so that the welder willhave convenient access for adjusting the wire feed speed.

Turn now to FIG. 9, which is a schematic diagram of circuit powersupplies 38. Input power is provided to power supplies 38 by +80 voltconductor 21 and 80 volt return conductor 23 which also serves as thecircuit ground. Regulator 453 and filter capacitors 455 and 456 provideregulated 14 volts to PIN 15 of modulator 457. Modulator 457 is aregulating pulsewidth modulator such as the SG3524B manufactured bySilicon General, Garden Grove, Calif. The SG3524B, although almostidentical in overall function to the SG3524/LM3524, is an improveddevice with additional protection features and should be used, incircuit power supplies 38, instead of the SG3524 manufactured by SiliconGeneral, or the LM3524 manufactured by National SemiconductorCorporation, Santa Clara, Calif. Capacitor 462 provides additionalfiltering for the +5 volt regulated output (PIN 16) of modulator 457.

Resistor 470 and capacitor 473 were selected to yield a frequency ofapproximately 15 kHz for the oscillator in modulator 457. This frequencyis not critical but should be within the input frequency specificationof power transformer 492. Resistors 463 and 465 form a voltage dividerwhich places approximately 2.5 volts on the non-inverting input (PIN 2)of the error amplifier of modulator 457. The inverting input (PIN 1) ofthis error amplifier is connected to circuit ground 23 through an inputcurrent balancing resistor 467.

The compensation input (PIN 9) of modulator 457 is connected to circuitground 23 through one microfarad capacitor 474. It will be appreciatedthat, as configured, and except as described below, the error amplifierof modulator 457 will charge capacitor 474 to approximately 4 to 5volts, thereby yielding an output pulsewidth of about 90 percent of theperiod of the frequency of the oscillator of modulator 457.

The shutdown input (PIN 10) of modulator 457 is not used and isconnected to circuit ground 23.

Overcurrent protection for transistors 477 and 484 is provided by thecurrent limiter of modulator 457. The CL- input (PIN 5) of modulator 457is connected to circuit ground 23. The CL+ input (PIN 4) of modulator457 is, as explained below, connected to sense the current passingthrough transistors 477 and 484.

If this current exceeds the safe value for transistors 477 and 484, thecurrent limiter of modulator 457 begins discharging capacitor 474 andlowering the voltage on the compensation input of modulator 457. Thisreduces the pulsewidth, and therefore the current through transistors477 and 484.

The collectors (PINS 12 and 13) of the output transistors of modulator457 are connected to +14 volt conductor 454 through 110 ohm currentlimiting resistor 460. The emitter (PIN 14) of the Q output transistorof modulator 457 is connected by conductor 475 to the base of NPNDarlington power transistor 477, one end of 47 ohm base shunt resistor480, and one contact of SPDT switch 481. The emitter (PIN 11) of thenegated Q output transistor of modulator 457 is connected by conductor476 to the base of NPN Darlington power transistor 484, one end of 47ohm base shunt resistor 483, and the other contact of switch 481.

The emitter of transistor 477 is connected to the anode of blockingdiode 468. The collector of transistor 477 is connected by conductor 490to the cathode of reverse voltage protection diode 469 and one end ofthe primary of power transformer 492.

The emitter of transister 484 is connected to the anode of blockingdiode 478. The collector of transistor 484 is connected by conductor 491to the cathode of reverse voltage protection diode 479 and to the otherend of the primary of power transformer 492. The center tap of theprimary of power transformer 492 is connected to +80 volt conductor 21.

The pole of switch 481 is connected to one end of 250 ohm potentiometer482. The wiper and other end of potentiometer 482, the other end of baseshunt resistors 480 and 483, the cathodes of blocking diodes 468 and478, the anodes of reverse voltage protection diodes 469 and 479, andthe CL+ input of modulator 457 are connected by conductor 487 to one endof 0.04 ohm current sensing resistor 488. The other end of resistor 488is connected to 80 v return conductor 23.

Ignoring, for a moment, switch 481, potentiometer 482, resistors 480 and483, and diodes 468, 469, 478 and 479, it will be appreciated thatmodulator 457, transistors 477 and 484, resistor 488, and transformer492 form a conventional inverter power supply. Current passing throughtransistors 477 and 484 passes through resistor 488 and the resultingvoltage developed across resistor 488 is applied to the CL+ input ofmodulator 457. The width of the output pulses of modulator 457 istherefore reduced if the current exceeds the desired value.

However, in a conventional inverter, transistors 477 and 484 must be amatched pair. If the match is not good, then one of the transistors willdraw significantly more current, heat up, draw even more current, and soon, with thermal runaway and destruction of the transistor being theusual result, frequently followed by destruction of the othertransistor.

Switch 481 and potentiometer 482 allow the safe use of unmatchedtransistors for transistors 477 and 484. Assume that transistor 477 hasthe higher beta and draws more current than transistor 484. Switch 481is therefore placed in the first position so that the base of transistor477 is shunted by potentiometer 482. Potentiometer 482 is then adjustedto absorb some of the output of modulator 457 so that transistor 477receives less base drive current and therefore draws the same collectorcurrent as transistor 484. The transistors 477 and 484 are now"balanced."

If transistor 484 has the higher beta, switch 481 would be placed in thesecond position and potentiometer 482 adjusted to reduce the base andcollector currents of transistor 484. It will be appreciated thatresistor 460 limits the available output current of modulator 457 sothat potentiometer 482 can draw base drive current away from transistor477 or 484, as appropriate.

The value of potentiometer 482 is not critical but should be largeenough that, when in the maximum resistance setting, negligible drivecurrent is shunted away from the transistor base, and low enough so thatthe adjustment is not confined to a very small portion of the range ofpotentiometer 482.

It will be appreciated that a prior art method of balancing unmatchedtransistors is to connect a potentiometer between the bases oftransistors 477 and 484 and connect the wiper of the potentiometer toconductor 487. However, this prior art method is unsatisfactory becauseof two problems. The capacitance across the potentiometer couples thebase drive signal from the base of transistor 477 to the base oftransistor 484, and vice-versa, thus simultaneously turning on bothtransistors 477 and 484, an undesirable, inefficient, and oftendestructive result. Also, there is a small, but significant, resistancebetween the resistance element and the wiper of the potentiometer. Thisalso allows cross-coupling of the base drive signals and can lead to thesame undesired, inefficient, and destructive result. In the preferredembodiment, potentiometer 482 does not connect between the bases oftransistors 477 and 484 and therefore there is no cross-coupling of thebase drive signals.

Assume now that switch 481 and potentiometer 482 have been set so thattransistors 477 and 484 are "balanced." Assume now that, because ofdifferent collector-to-case heat transfer characteristics, differentcase-to-heatsink heat transfer characteristics, or some other reason,transistor 477 becomes hotter than transistor 484. Transistor 477 willthen draw more collector current, which causes it to become even hotter,and so on, with thermal runaway and destruction again being the likelyresult.

Resistors 480 and 483 act to prevent this undesired event fromoccurring. It will be appreciated that resistors 480 and 483 shunt basedrive current away from transistors 477 and 484, respectively. Resistors480 and 483 have a positive resistance-temperature coefficient. Resistor480 is thermally connected 486 to the case of transistor 484. Resistor483 is thermally connected 485 to the case of transistor 477. Iftransistor 477 draws more collector current and becomes hotter thannormal, it will heat up resistor 483 hotter than normal. This increasesthe resistance of resistor 483 so transistor 484 receives more basedrive current and draws more collector current so that transistors 477and 484 remain "balanced." Also, the increase in current causesmodulator 457 to decrease the output pulse width, thereby limiting thetotal average current of transistors 477 and 484 to a safe value.

In the preferred embodiment, transistors 477 and 484 are 350 volt, 20amp, NPN Darlington power transistors, such as the MJ10000, manufacturedby Motorola, Inc., Phoenix, Ariz. The MJ10000 has an internal diode,with its cathode connected to the collector, and its anode connected tothe emitter, to help prevent a negative collector voltage fromdestroying the transistor. However, the inventor has found that, in thepreferred embodiment, the negative collector voltage caused bytransformer 492 did destroy transistors 477 and 484. It is not knownwhether the failures occurred because the internal diode of the MJ10000was unable to handle the current, the base-collector junction becameforward biased and the MJ10000 operated in the inverted mode, or becauseof some other mechanism. The use of higher power, more expensive powertransistors for 477 and 484 did solve the problem but increased thecost.

Diodes 468, 469, 478 and 479 prevent these failures from occurring.Diodes 468 and 478 prevent transistors 477 and 484 from operating in theinverted mode and also prevent current from passing through theirinternal diodes. Diodes 469 and 479 are then necessary to absorb thenegative voltage spikes produced by transformer 492. Diodes 469 and 479should be fast recovery diodes. This allows the use of the lower power,less expensive MJ10000's for transistors 477 and 484.

Transformer 492 has several secondary windings. One of these windings isconnected to 14 volt rectifier & filter 495. Rectifier & filter 495provides +14 volts on conductor 104 and -14 volts on conductor 106. The14 v return is conductor 105. A second winding is connected to 7 voltrectifier & filter 494. Rectifier & filter 494 provides +7 volts onconductor 100 and -7 volts on conductor 102. The 7 v return is conductor101. A third winding is connected to 15 volt rectifier & filter 493.Rectifier & filter 493 provides a regulated +15 volts on conductor 440.The 15 v return is conductor 441. Separate secondary windings andseparate return conductors are used because the circuits that arepowered operate at different circuit "ground" potentials. The design ofrectifier & filter 493, 494 and 495 is well known to those skilled inthe art.

The +80 volt conductor 21 is connected to the positive terminal offilter capacitors 500 and 502, one terminal of high frequency filtercapacitor 503, and to the "ground" input of -13.3 volt voltage regulator501. The -13.3 volt output of regulator 501 is connected to the otherend of capacitors 502 and 503 and provides a regulated -13.3 volts onconductor 430. The voltage on conductor 430 is a regulated -13.3 voltwith respect to +80 v conductor 21. Conductor 430 therefore has anunregulated, nominal potential of +66.7 volts with respect to 80 vreturn conductor 23.

The V_(EE) input of regulator 497 is connected to the anode of blockingdiode 497. The cathode of diode 497 and the other end of capacitor 500are connected by conductor 431 to one end of 10 ohm resistor 496. Theother end of resistor 496 is connected to 80 v return conductor 23.Conductor 431 provides the unregulated +1 volt signal which is used inpulsewidth modulators 32 and 34 of FIG. 2. The total current throughregulator 501 is approximately 100 milliamps and develops the 1 voltdrop across resistor 496.

The preferred embodiment of the present invention discloses a weldingsystem with a central welding power supply and numerous, high efficiencyweld selector stations and also discloses a weld selector station withthe capability of adjusting arc characteristics, with automatic inputvoltage compensation, with the capability of accommodating differenttypes of welding, and with a power supply which utilizes inexpensive,unmatched, low power transistors. Although the preferred embodiment ofthe present invention has been described with particularity, it will beunderstood that numerous modifications and variations are possible.Accordingly, the scope of the present invention is to be limited only bythe claims below.

I claim:
 1. An electronic welder, comprising:a power supply for providing operating voltage and operating current; a voltage divider responsive to said operating voltage for providing a first signal; control means for providing a second signal; a pulsewidth modulator for providing a third signal having a pulsewidth responsive to said first signal and said second signal; and output means responsive to said third signal for providing welding power to a welding operation.
 2. The electronic welder of claim 1 wherein:said voltage divider comprises a first resistor, a second resistor, and a third resistor, connected in series, said second resistor having a resistance significantly less than the resistance of said voltage divider; and said first signal is the voltage developed across said second resistor.
 3. The electronic welder of claim 2 wherein:said pulsewidth modulator comprises an operational amplifier having a differential input; and said voltage across said second resistor is applied to said differential input.
 4. The electronic welder of claim 3 wherein said operational amplifier is a transconductance amplifier operating in an open loop configuration.
 5. The electronic welder of claim 4 and further comprising:a load connected to the output of said transconductance amplifier, said load having a resistance such that said transconductance amplifier operates in a linear region when said voltage across said second resistor is applied to said differential input.
 6. The electronic welder of claim 1 wherein said pulsewidth modulator decreases said pulsewidth in response to an increase in said operating voltage.
 7. The electronic welder of claim 1 and further comprising:a voltage regulator for providing a regulated voltage; and means connected to said power supply and responsive to said operating voltage for providing an unregulated voltage; wherein said voltage divider is connected between said regulated voltage and said unregulated voltage.
 8. The electronic welder of claim 7 wherein said pulsewidth modulator comprises said voltage regulator.
 9. An electronic welder for use with an external power supply, comprising:input terminals for receiving operating voltage and operating current from said external power supply; a voltage divider responsive to said operating voltage for providing a first signal; control means for providing a second signal; a pulsewidth modulator for providing a third signal having a pulsewidth responsive to said first signal and said second signal; and output means responsive to said third signal for providing welding power to a welding operation.
 10. The electronic welder of claim 9 wherein:said voltage divider comprises a first resistor, a second resistor, and a third resistor, connected in series, said second resistor having a resistance significantly less than the resistance of said voltage divider; and said first signal is the voltage developed across said second resistor.
 11. The electronic welder of claim 10 wherein:said pulsewidth modulator comprises an operational amplifier having a differential input; and said voltage across said second resistor is applied to said differential input.
 12. The electronic welder of claim 11 wherein said operational amplifier is a transconductance amplifier operating in an open loop configuration.
 13. The electronic welder of claim 12 and further comprising:a load connected to the output of said transconductance amplifier, said load having a resistance such that said transconductance amplifier operates in a linear region when said voltage across said second resistor is applied to said differential input.
 14. The electronic welder of claim 9 wherein said pulsewidth modulator decreases said pulsewidth in response to an increase in said operating voltage.
 15. The electronic welder of claim 9 and further comprising:a voltage regulator for providing a regulated voltage; and means connected to said power supply and responsive to said operating voltage for providing an unregulated voltage.
 16. The electronic welder of claim 15 wherein said pulsewidth modulator comprise said voltage regulator.
 17. The electronic welder of claim 16 wherein said unregulated voltage is derived from a predetermined one of said input terminals.
 18. For use with an electronic welder which provides a pulsed output having a selectable pulsewidth, a method for improving the performance of said electronic welder, comprising the steps of:monitoring an input voltage of said electronic welder; and changing said pulsewidth in response to changes in said input voltage.
 19. The method of claim 18 wherein said step of changing comprises decreasing said pulsewidth in response to an increase in said input voltage.
 20. The method of claim 19 wherein said step of monitoring comprises measuring the voltage across a predetermined resistor in a voltage divider connected to said input voltage, said predetermined resistor having a resistance which is significantly less than the resistance of said voltage divider. 